Fullscreen HTML5Med Res (??MB)HTML4

AUDIO OUT AUDIO OUT L R By Jake Rothman Discrete audio op amp – Part 2 L ast month, I introduced a Current mirror Note: resistors and capacitors have been renumbered from Fig.9 and Fig.12 Special compensation network R3 330Ω Output stage bias C1 220nF R19 1kΩ OA+ Noninverting input C8 150pF TR5 BC556 1mA R1 180Ω + R6 2.2kΩ C2 1µF 10V + R19* 1kΩ R8 220Ω OAO R17 C6 47kΩ 1µF + 35V + C10 100µF 25V Output C15 100pF Mute (Pull to 0V or V–) R25 560Ω or link TR14 BD139 V– –6 to –25V OAO High-power option For low-power operation R24/R25 are links for high-power operation use 560Ω resistors C10 100µF + + 0V + R23 100kΩ R15 43kΩ R16 5.6kΩ VAS current sink Output – V– C8* 150pF R22 47Ω 0V Output isolator R18 22kΩ R12 12Ω TR10 BC556 L1** 3.9 to 10µH V+ OA– C5 100nF C11 10pF Feedback components OA+ TR11*a BC556 R14 1.5kΩ R21 22kΩ Discrete op amp C7 4.7µF 4.5mA TR12 BC546 R20 4.3kΩ R11 12Ω Thermal link TR7 BC546 LED red low I Output stage L1 70mV 3.9 to 10µH R13 2.7kΩ Iq set PR2 5kΩ C9 100µF 10V Current sink voltage reference TR8 BC546 R7 10kΩ + LTP current sink R10 3.9kΩ 0V OA– Inverting input C4 10µF + 10V R9 6.8kΩ C13 680pF R2 180Ω R5 470Ω TR13 BD140 6mA TR3 BC556 C12 R26 47pF 3.3kΩ 1mA TR6 BC546 Long-tailed pair R24 560Ω or link TR9 BC546 TR1 TR2 BC546 BC546 R18 22kΩ 46 V+ High-power option +6 to +25V (symmetrical) Short-circuit protection PR1 5kΩ TR4 BC556 C14 + C7 4.7µF 10µF 25V C3 39pF (6-10 gain) 82pF (unity gain) R4 330Ω DC Offset Input VAS R27 330Ω Input + We are so used to thinking of op amps as little black boxes that we rarely consider what we could achieve by tweaking component types or values month, I want to further explain my thoughts on this circuit, especially the many options open to builders thanks to its non-integrated format. novel project – my design for a discrete op amp (DOA) primarily intended for audio applications. This R21 R20 22kΩ 4.3kΩ *RF filtering on input (gain set x6) **Isolation from cable capacitance C9 100µF (non-polar preferred) R23 100kΩ For high-power operation R11/R12 are 2.2Ω For high-power operation fit C15 if using BD139/140 Fig.15. The DOA circuit with each stage defined by blocks. (Note, a few gremlins crept into last month’s diagram of the overload part of the circuit in Fig.13. TR11 had its base and collector flipped; C6 was reverse polarised and must go to ground not V–. Also R17 should be 47kΩ, not 4.7kΩ.) Practical Electronics | October | 2023 Discrete op amp V+ Input OA+ + OA– – OAO Output V– R21 R20 22kΩ 4.3kΩ (gain set x6) + C9 100µF Gain = 1 + (R21/R22) Divide R21, R22 by 10 for lower noise 0V Fig.17. DOA configuration of non-inverting amp. n S erviceable n U pgrade possibilities – there are endless tweaking and experimenting opportunities n G ood-quality discrete PNP transistors are available, whereas Fig.16. PCB for the DOA – we’ll construct it next month. – this is the great opportunity offered by a discrete version of the classic op amp. Figure 13 update The basic DOA circuit diagram was given in Fig.13 last month. In fact, there are a number of components to be added to enable various amplifiers (eg, inverting or non-inverting) to be built around the op amp on the PCB. The complete circuit is shown in Fig.15. There are specific inputs and outputs for the DOA (OA–, OA+ and OAO), and also for complete amplifiers with added feedback and coupling components. Each of the DOA’s internal stages are highlighted in Fig.15. These are the: n L ong-tailed pair difference amplifier (LTP) n L TP current mirror load n L TP current sink n V oltage reference for LTP current sink n V oltage amplifier stage (VAS) n V AS constant current (sink) n O utput stage n O utput bias stage n H igh-power output stage n S hort-circuit protection system A quick note about the short-circuit protection system, which I didn’t cover last month. This operates by sensing the voltage between the emitter (TR9 and TR10) resistors R11 and R12. When the voltage across TR11’s base-emitter junction exceeds 0.6V, it turns on. This pulls the base of TR12 towards 0V turning it on, thereby shutting down the DOA. Winners and losers Before going any further I’d like to summarise the pros and cons of taking the discrete route, since it played a big role in how I designed the circuit. Pros n L ower noise if special input transistors are used n H igher possible power rail voltages, giving more headroom n L ower distortion into a low impedance load (<600Ω) – the result can effectively be a small power amplifier most chip designs only have low-gain lateral PNPs unless expensive proprietary processes are used n C urrents and voltages can be adjusted and optimised to suit the application n L ow-cost, commodity parts can be used – some special chips cannot be imported into some locations/regions n ‘ Discrete’ has fashionable connotations in high-end audio – potentially higher profits! n L arge, high-quality NP0 capacitors can be used for compensation. On the other hand, ICs can only use poor-quality oxide-dielectric capacitors. In ICs the maximum capacitance value available is greatly reduced by the (relatively) huge amount of chip real estate consumed by capacitors n D iscrete designs are fun to build and highly educational for the home constructor n Y ou can put your name on it via the silk screen! Cons n H igh parts count n P oorer reliability compared to ICs n H igh labour cost, unless you build it yourself n L arge size, so not a good idea for systems using half a dozen op amps. The basic PCB is shown in Fig.16. n L ower levels of protection, more vulnerable to overheating and rail-to-output shorts n H igher power consumption n L ess consistent unit-to-unit performance n H igher offset and input bias currents, so generally less suitable for precision DC instrumentation Configuration options One of the nice advantages of a PCB-based DOA design is that we can include on the same board space for standard op amp circuit ‘external’ components – for example, input and feedback resistors. This means on one board we can easily create some of the most common op amp circuits. Non-inverting amplifier This will possibly be the most popular option used by most audiophiles for pre-amps and headphone amplifiers. It is ideal as a pre-amp following a volume control. The configuration of the non-inverting op amp is shown in Fig.17. A finished PCB is shown in Fig.18. n L ower noise when being driven by a low source impedance Inverting amplifier – eg, moving coil transducers n L ower cost than specialist audio ICs The inverting amplifier configuration is shown in Fig.19. This gives lower distortion than the non-inverting amplifier but at Practical Electronics | October | 2023 47 R21 All resistors same value, eg 1kΩ V+ Omit C9, R18 R20 Input– OA– OA+ Input+ – Output + OAO R19 This resistor in C8 position V– Discrete op amp 0V Fig.20. DOA configuration of differential amplifier. Fig.18. Finished PCB for low-power DOA. Note unused extra output transistor holes. R21 V+ R20 OA– Input – OA+ Omit C7, C9 Link R18 and R19 R20 beomes Rin Output + Gain = –R21/R20 OAO V– Discrete op amp Fig.22. (above) Distortion plot of DOA with a load of 600Ω, ±25V PSU and 12Vpk-pk output. (This includes noise.) 0V Fig.19. (left) DOA configuration of inverting amplifier. the expense of higher noise from the input resistor R20. The inverting configuration is useful for mixing and distortion-cancelling negative output impedance transformer drivers (see Audio Out, February 2022). Differential amplifier The configuration for this is shown in Fig 20. It’s useful for interfacing with low-impedance sources, such as moving-coil microphones. Buffer To be honest, this is just included for ‘completeness’ and I do not recommend using it! There’s not much point using a full DOA as a follower, since the massive open-loop gain of an op amp is not needed. A simpler circuit using several transistors could be used, such as a diamond buffer, which can give Discrete op amp V+ Input– Input+ OA– + OA+ – OAO Output V– 0V Link R21 Omit C9, C11 and R20 Set C3 to 82pF and C13 to 1nF Fig.21 DOA configuration of buffer amplifier. 48 0.001% total harmonic distortion (THD). If you really want to make one then the configuration is shown in Fig.21. Discrete op amp specifications Now that we have our DOA design, what is its performance like? Table 1 below shows a mini data sheet for the basic DOA using the BC546B and BC556 transistors. This is a preliminary specification – more refined data will emerge after I’ve tested multiple PCBs with different devices. Transistor options Many transistors are available with different Hfe grades, or put another way – ordinary, good and excellent versions. Table 1. Discrete op amp ‘data sheet’ Max supply voltage.............................±30V (recommended ±25V) Output swing total..............................power rail minus 1.4V loss with no load Total current consumption.................12-18mA quiescent (output Iq set by user) Max current, clipping into 330Ω........40mA Absolute max current output.............±100mA limit = 60mA Input bias current...............................2.7µA Noise voltage.......................................1.5nV/√Hz with low-noise input transistors Current noise......................................1.2pA/√Hz with low-noise input transistors Optimum source impedance..............1kΩ Offset voltage, un-trimmed.................±150mV, trimmed ±2mV Using dual transistors (TR1/2/4/5).....<±10mV untrimmed Max dissipation..................................600mW Continuous power output...................300mW into 330Ω Open loop gain....................................>100,000x (no emitter resistors, R1 and R2) Gain bandwidth product....................13MHz Slew rate.............................................10V/µS proportional to transistor currents Input impedance.................................>1MΩ Output impedance (open loop)..........20Ω Min load impedance...........................330Ω (This can be reduced by a factor of 10 by adding extra output transistors.) Power supply rejection ratio PSRR....−80dB negative rail −74dB positive rail (no decoupling network) Distortion............................................<0.002% THD at 20Hz-20kHz into 600Ω (see Fig.22) Practical Electronics | October | 2023 Toshiba dual transistors 6 6 5 4 D 1 G Q1 1 2 3 HN1A01F PNP 5 4 Top view C E B C E BC546 BC556 BC549 BC550 BC212 Q2 B Japanese pin out 2SA970BL BC212L (L suffix for centre collector) Type name 1 2 3 6 5 4 hFE rank G (best) = 200 to 400 6 5 4 C 1 Y 1 2 3 Instaling ‘Japanese’ transistors in PCBs (top view) B C E C Rotate 180° B ‘Cross legs’ Use sleeving E Type name HN1C01F NPN Fig.24. The output transistors and bias transistor are all thermally coupled together on a metal pad on the PCB. Q2 Q1 1 2 3 tary NPN/PNP pair. High current and voltage capability are the main requirements, along with reasonable Hfe (>200). BC546 and BC556 are only rated at 100mA. BC639 and BC640 are rated at 80V 600mA, but their low Hfe of 80-120 means there may be a need to increase the VAS current. ZTX651 and ZTX751 are similar alternatives, rated at 65V 2A. hFE rank Y = 120 to 240 Fig.23. Transistor packs and pin outs, and how to install centre-collector TO92 devices. For audio work, the higher Hfe values are preferred since they provide lower distortion. On graded BC series transistors (eg, BC549) the suffix ‘A’ is for the lowest Hfe values of 100-200; for ‘B’, 200-350; and for C, >350-600. If no grade is quoted then the gain can vary from about 100 to 600. On Japanese Toshiba leaded small-signal devices , such as the 2SA970, ‘BL’ denotes the highest H fe group. Unfortunately there seems to be a multitude of suffixes and consulting the data sheet is the only option. Toshiba, Hitachi and Sanyo generally make the best audio transistors if you can get them, but the circuit still gives excellent results with generic European types. Note that most Japanese transistors have their collector leads in the middle rather than the more common centre base, as shown in Fig.23. Once I’ve built several PCBs with different devices I’ll have a better feel for the what the real benefits of Japanese transistors are. The component option list below is in order of quality – in other words, upgrading the LTP circuit has the most beneficial effect. For a small current mirror noise improvement, the BC550 has specified lower noise at 1-3dB NF, but a lower 45V Vce (only an issue for high-power rails) The 2SA2362K is better still with 0.5dB NF. Output stage bias generator In this circuit for TR8, it’s best to use the same NPN device as in the output pair (TR9). This will give better thermal tracking since their V be values will be similar. So, if you are using ZTX output devices, use a ZTX651 for TR8. Note how the output transistors TR9 and TR10 are all thermally coupled together with the bias transistor TR8, as shown in Fig.24. The remaining transistors (TR8/11/12) are uncritical (low voltage, low current, low Hfe). Old timers like the NPN BC108 and BC182 can be used for TR6 and Dual devices Dual devices are worth considering because they automatically come with well-matched transistors. I was pleased to discover these dual devices from Toshiba: HN1C01F-GR NPN for TR1/ TR2, and HN1A01F-GR PNP for TR4/ TR5. They can be mounted either way round (see Fig.23) which makes soldering a little less error prone. They are available from Mouser in SOT26-6 6-pin packs, which are ‘easy’ to solder. Last, but not least, they are cheap. VAS For TR1 and TR2, low noise and matching is critical. BC546B is recommended for normal use. The BC549C (max rail ±25V) will give lower noise. For best results use 2SC2240BL, which have a very low 0.5dB noise factor (NF). So long as the rail voltages do not exceed ±18V, a good choice would be the 2SC3068, which has an Hfe of 1600. This device is almost a possible substitute for JFETs, having a quarter of the normal bipolar transistor input bias current. The VAS PNP transistor TR3 is subjected to the full rail voltage, so it will need to be rated at a minimum Vce of 65V. It also needs a high Hfe to avoid loading the current mirror. Higher voltage types generally have lower Early effect distortion, but with lower Hfe, which then increases the loading on the current mirror. Ultra-low noise is less important, since it mainly amplifies the first stage noise. A BC556 is acceptable, but the BC556B has higher Hfe. The Japanese 2SA2362K device is rated at 120V with an Hfe of 350. The associated current sink transistor (TR7) needs a high voltage rating, but its Hfe and noise specifications are not important. A BC546 is fine here. Current mirror Output stage In a standard DOA current mirror, use BC556 PNP transistors for TR4 and TR5. For the low-power output DOA, TR9 and TR10 are a complemen- Fig.25. Installing FETs on the input LTP stage. Long-tailed pair (LTP) Practical Electronics | October | 2023 R3 330Ω R4 330Ω DC Offset PR1 5kΩ TR4 BC556 C1 220nF TR5 BC556 1mA 1mA TR1 BF244A Non-inverting input Top view Inverting input TR2 BF244A Replace R1, R2 with links R6 2.2kΩ TR6 BC546 D/S G S/D R5 470Ω + C2 1µF 10V Use centre pin gate (G) device such as BF244A Note JFET drain (D) and source (S) are interchangeable 49 TR11. BC178 and BC212 could be used for PNP TR12. Audio amplifier with 10kHz input Amplifier output Slight fuzziness (low-level oscillation) Amplifier output Audio amplifier with 10kHz input Oscillation burst often at one particular level Square wave 10kHz input Settling time Output waveform Overshoot Fig.26. Amplifier on the verge of oscillation. Input sinewave 20kHz Output slewing-induced distortion Large voltage swing FET input The most important possible tweak/upgrade to the standard DOA (Fig.15) is using FETs for the LTP input transistors (TR1/2) shown in Fig.25. It’s worthwhile substituting N-channel JFETs here to produce high input impedance (>1MΩ) and low bias currents. The gate becomes the base connection and since JFETs are symmetrical devices, it does not matter which way the source and drain pins go into the emitter and collector holes. The voltage noise is twice as much for generic JFETs compared to bipolar transistors, but the current noise and bias currents are very low. It’s more difficult to match JFETs, since their spreads are much higher than bipolar transistors. The distortion is also higher, although using BF244A FETs on a ±25V supply the difference becomes negligible. FET inputs are particularly good for active filters and circuits using potentiometers without coupling capacitors, such as amplifiers following high-impedance volume controls. The 2SK170 is the best device. Older European low-noise JFETs, such as the BFW10 are also good. If you are rich, consider dual matched JFETs from InterFET. High power output option Ramps from capacitor charging and discharging. Often asymmetric due to different current in postitive and negative direction Fig.27. Effect of overloading the input stage by too large a value for C3. Standard ‘Miller’ compensation C3 39pF VAS C12 82pF C3* 82pF 2-pole compensation Vary resistor from 100kΩ (no filtering effect) to 2kΩ (maximum filtering); typical value is 3.3kΩ. R26 0V Tune for minimum distortion at 10kHz without oscillation. VAS *For C3 connection details see next month Fig.28. Second-order compensation. 50 A pair of extra output power transistors (TR13 and TR14) can be added to the existing output stage on the PCB, which then becomes the driver stage. This enables more difficult loads to be driven down to 50Ω. This will be covered next month. Stability tip Remember that output stages can oscillate with capacitive loads, such as screened audio leads and scope probes. This is normally compensated for by inserting a resistor (39Ω to 120Ω) in the output (R22). This unwanted additional output impedance can be mitigated by putting an inductor (L1) in parallel with the resistor. Another stability tip involves adding a small ‘phase lead’ capacitor of 4.7 to 47pF (C11). This is often placed across the feedback resistor (R21), and compensates for any stray capacitance to ground from the inverting input. It is best optimised by looking for any overshoot on a high frequency (>10kHz) square wave. Too much capacitance will cause low-pass filtering and round-off the square wave. Compensation options In all amplifiers using negative feedback, a capacitor (Cdom, C3 on the DOA board) is needed to stabilise the loop by setting a dominant high-frequency roll-off. The more feedback, or lower the closed-loop gain, the bigger the capacitor. The single compensation capacitor C3 has to be 82pF for a unity gain (buffer), such as in a Baxandall tone control, but 39pF for a gain of 6-10 and scaled accordingly: Gain 1 (unity) 6-10 10-20 20-50 C3 82pF 39pF 27pF 15pF Do remember, these are guidelines only, you still need to check stability for your board, application and batch of transistors. C3 can be optimised for different gains and transistor spreads by using a 5.5 to 65pF trimmer capacitor instead of C3. Remember to increase it by at least 30% beyond the point where oscillation stops, just in case changes in temperature and loading cause more compensation to be needed. Instability often occurs at only some points in a sinewave, as shown in Fig.26. Stability compensation includes so many unknown variables it often becomes a fiddle fest. This is why you see odd ceramic capacitors soldered under the board in many commercial amplifiers. Another thing that has to be watched is that this capacitor does not get too big (over compensation), because it may then overload the input stage. There has to be enough current available to cleanly charge or discharge it on rapidly changing signals. This can be tested by driving the amplifier to almost full output voltage at 20kHz and is revealed if the output sinewave suffers from slew limiting and becomes an asymmetrical ramp wave, as shown in Fig.27. Second-order compensation If you want an amplifier with the lowest overall high-frequency distortion, then to achieve more overall feedback at high frequencies, C3 is upgraded to a second-order compensation network. This increases the slope of the compensation curve, so it can be moved to a higher frequency before oscillation occurs, allowing the feedback factor to be higher at higher frequencies. This gives lower distortion at high frequencies. It was used in the Tiny Tim Amplifier (PE, January 2015). It normally adds another capacitor in series with Cdom (C3) with the junction tied to earth via a resistor to Practical Electronics | October | 2023 V+ 10 to 30V High-pass filter Second output inclusive compensation path C12 47pF C3 (Cdom) 33pF 10kΩ R26 3.3kΩ 5.6V BZY88 0V + Input – VAS Output *May need to be adjusted Metal film 2.7kΩ C13 680pF 10µF+ 10V Tant Output 0V Fig.29. Inclusive Baxandall second-order compensation. You couldn’t do this on a chip. provide the extra high-pass slope. This scheme is shown in Fig.28. Inclusive Baxandall secondorder compensation Normally, in amplifiers the compensation capacitor feedback loop cannot include the output transistors. This is because large transistors, such as the TIP31, are rather slow, often with a transition (unity-gain) frequency (Ft) of only 3MHz. This phase lag from the output transistors can cause oscillation. In DOAs, the smaller output devices used usually have an Ft at least 20x higher, allowing the compensation loop to include the output devices. This enables crossover distortion to be more effectively reduced by the overall negative feedback, minimising the distortion rise often seen around 10kHz. It also seemed a good idea to combine this with second-order compensation. This compensation scheme is shown in Fig.29 and it can be seen that a signal from the output is fed into C13, rather than from the output of the VAS. This signal is high-pass filtered by C12 and R26 to get the second-order slope. Since there is now feedback summed from two points, the VAS feedback can be reduced by decreasing C3 to 33pF. This reduces the loading on TR1. The overall effect is to provide inclusive Baxandall second-order compensation. Note that for gains other than 10, the values of C3, C12 and C13 will all have to be scaled; more capacitance for less gain. I will cover this in more detail next month. Remember to only use low distortion dielectric capacitors such as NP0 ceramic, polypropylene or polystyrene for C3, C12 and C13. If you use low voltage SMT X7R ceramic types, they cause the distortion we’re trying to reduce. Optional off-board input bias current compensation For a bipolar transistor version of the DOA, bias current flows into the base of the LTP input transistors (DOA input terminals). This causes a small negative offset which Practical Electronics | October | 2023 2.7MΩ* Bias current 2.1µA Discrete op amp 2.7MΩ* V+ OA+ + OA– – OAO V– Fig.30. Input current bias compensation – take care with noise injection. is why the electrolytic capacitors (C7, C9 and C10) in Fig.15 are all oriented with the negative terminal facing into the DOA. If you have an application that requires cancellation of these input bias currents then you can add on the circuit shown in Fig.30 (an offboard add-on). It works by injecting a positive current from a low-noise source. With PNP transistors the current has to go the other way. Off-set adjustment The voltage input offset trimmer (PR1) can either null the output (OAO) offset or balance the LTP collector (TR1/2) currents for minimum distortion. It can’t optimise both at the same time. Sonically, even-order distortion is much more subtle than the effect of earlier asymmetrical clipping, so trim for minimum OAO voltage offset. It should be possible to trim this to within ±2mV. Quiescent current adjustment The output stage quiescent current (Iq) must be set carefully with PR1. If set too high the output transistors will burn out. Always set for maximum resistance (fully clockwise) before turning on. Gradually increase until 70mV is obtained across each 12Ω emitter resistor (R11 and R12) giving 6mA. This current can be optimised for lowest distortion and current consumption for various loads. More details will be provided next month. Input and output protection formers and loudspeakers. This protection is shown in Fig.31. Normally a DOA is built into a system and doesn’t need these diodes, but in the less common situation where you are directly driving an inductive load or are using phantom power then consider adding them. Recommended power supply I view an amplifier as a power supply modulator. If the power supply is poor quality, then so will be the amplifier’s output. Unlike power amps, a regulated supply is always used for op amps, so the voltage ratings of the transistors are less critical because the rails are better specified. For the DOA I recommend my special low-noise audio power supply described in Audio Out, April 2022. A suitably scaled-up version providing ±25V is shown in Fig.32. Note that to set the regulators to 25V, unusual resistor values (R7 and R8) are needed. E96 series 1% metal-film resistors are now cheap and are available from Mouser and Farnell. If you want, 2.2kΩ 0.5W and 82Ω resistors can be connected in series which will come closer to 25V than the E96 resistors. All amplifiers have a power supply rejection ratio (PSRR), that is how much noise on the power rail is attenuated at the output. This parameter is normally specified in dB at a particular frequency. In this case, it is around −75dB at 1kHz. With this design, the PSRR is 6dB worse on the positive rail because signals from the rail can be directly injected into the emitter of TR3. This is why an additional power supply decoupling capacitor is provided: C14 on the DOA. Another optional off-board add-on is backto-back diodes across the DOA inputs to protect TR1 V+ and TR2 from reverse biasing of their base-emitter juncD1, D2 D3 1N4148 OA+ 1N4001 + tions. This can permanently damage them, increasing Inputs Output OA– – OAO noise levels. Also, reverse-biD4 ased diodes from output to 1N4001 V– rails will protect the output Discrete V– op amp transistors from reverse voltage spikes from inductive loads such as output trans- Fig.31. Protection of inputs and outputs from voltage spikes. 51 L F1 1A A/S N Fig.32. New values for ±25V power supply using LM317/337 regulators. Obtaining a 22-0-22V transformer may be tricky. Cricklewood electronics do a big 80VA one and RS and CPC do smaller ones. Using the more common 25-0-25V type could exceed the 37V input voltage of the regulators and stress the 35V capacitors. T1 22-0-22V 7VA (minimum) S1 E IEC filtered mains connector 120V VDR1 275V 0V 22V 120V 22V 0V 0V 0V Con1 Screen R5 1Ω 2.5W +34V R1 10Ω C1 100nF D1 SB350 D3 SB350 R3 10Ω C5 1nF D5 UF4004 +34V Unreg Transformer connector Mains earth connect to metalwork C3 100nF R2 10Ω + C2 100nF Dirty Gnd star D4 SB350 R4 10Ω C4 100nF R13 1kΩ 0.5W C6 1nF C9 2200µF 35V + –34V –34V Unreg C17 10nF + C10 2200µF 35V + Clean Gnd + R7* 2.26kΩ C13 22µF 35V C14 + 22µF 35V R10 120Ω Adj –33V +25V R9 120Ω R8* + 2.26kΩ D8 UF4002 C12 100nF C8 2200µF 35V R6 1Ω 2.5W D7 UF4002 C11 100nF C7 2200µF 35V D2 SB350 IC1 In LM317 Out Adj +33V R11 1Ω + D20 1N4001 C15 22µF 35V C16 22µF 35V D21 1N4001 Con2 (x3) Op amp power R12 1Ω In IC2 Out LM337 Adj –25V *R7/R8 are Vishay MSR25 0.6W D6 UF4004 To work effectively this needs an associated decoupling resistor (R27). (On the early PCB shown in Fig.16, this was inadvertently omitted but is present in the ones supplied by PE.) Instrumentation use Instrumentation normally demands DC precision with low offset. DOAs have high unit-to-unit spreads, especially if they use random unmatched input transistors for TR1 and TR2. If the transistors are closely matched for Vbe at a low current and reasonably matched for Hfe, the DC performance will be adequate. Matching of the current mirror transistors TR4 and TR5 is less critical, but the Vbe values need to be close. If you don’t have access to a transistor analyser you can just put 3-pin sockets in the board and pick out pairs giving the lowest offsets and closest currents. The simplest way to do this is to solder in turned-pin socket strips, as shown in Fig.33. Mike Grindle (our PCB designer) has positioned the transistor pairs so they are thermally coupled with their flat sides facing each other to reduce drift. He has also provided pads for the dual SMT Toshiba devices (shown in Fig.34). Even untrimmed, these devices will give much better DC performance than separate unmatched transistors. JTAG Connector Plugs Directly into PCB!! No Header! No Brainer! Next month In Part 3 next month we will build the PCB and explore the higher power DOA option. Our patented range of Plug-of-Nails™ spring-pin cables plug directly into a tiny footprint of pads and locating holes in your PCB, eliminating the need for a mating header. Save Cost & Space on Every PCB!! Solutions for: PIC . dsPIC . ARM . MSP430 . Atmel . Generic JTAG . Altera Xilinx . BDM . C2000 . SPY-BI-WIRE . SPI / IIC . Altium Mini-HDMI . & More Fig.33. Trying different transistors is easy if you install sockets. In this case I was trying different types for TR3. 52 Fig.34. Toshiba dual transistors – note TR1 marking is ‘C1Y’ (for TR2 it is ‘D1G’). Solder these devices first. They can go in either way. www.PlugOfNails.com Tag-Connector footprints as small as 0.02 sq. inch (0.13 sq cm) Practical Electronics | October | 2023