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Create Fantastic Electronic systems using VERSION 10 10% off your first Flowcode purchase using code: EPE20 Use code at checkout: flowcode.co.uk/buy • NOW FREE FOR HOBBYISTS • The Fox Report Barry Fox’s technology column Project challenges for inventive PE readers T his month, I humbly offer creative readers two practical test problems which are crying out for DIY project solutions. The suggestions flow from my recent experience of rebuilding a home AV system, by replacing the main amplifier. This involved ripping out and re-connecting a jungle of wires, while simplifying the set-up by removing redundant components. Check cables Flat digital HDMI and Ethernet cables are now available. They hide neatly under rugs, but they love to twist themselves and treading on the twist can cause faults. A £10/20 battery-powered two-part continuity tester is an essential tool for checking if a digital cable has gone bad. Avoid the ‘touch’ test Rebuilding any AV system is a lot easier if you follow a few simple practical guidelines, not all of which will be as egg-suckingly obvious to others as they are to some. Some of these guidelines date back to invaluable tech training I received in the RAF. Optical SP/DIFs can be identified by looking for the telltale red laser light. Checking and identifying low-voltage audio cables can of course be done by touching to induce mains hum. But if the amplifier volume is up, there’s a risk of blowing speaker cones. Touching wires is always best avoided. You never know when a dangerous voltage may have crept through. Take your time – one labelled wire at a time Build a cable tester Wherever possible, disconnect only one wire or pair of wires at a time. Identically label each end of each wire. There are not enough colours in the rainbow to colour code every connection path. I use a Dymo label computer printer with Dymo software (but cheaper compatible label cartridges bought on line) to print simple stick-on labels for each end of each cable run. Connecting a meter to small phono and coax plugs is tricky and for years I have been happily using an analogue cable test kit from Vision Products of Northampton. This uses a low-voltage transmitter and receiver that plug into cable ends to show a red LED for short circuits and beep for successful connection. These handy testers came with an assorted collection of plugs and sockets that connect to almost every imaginable analogue cable. That’s the good news – the bad news it that Vision Products informed me that unfortunately these kits are no longer available, and I can’t find any other source. Perhaps someone would like to make this a construction project? It shouldn’t too difficult. A few simple rules... Use cable ties – gently Modern ‘handcuff’ cable ties are great for tidily binding cables together, to stop self-tangling. But don’t over tighten or you won’t be able to identify troublesome wires by gently tugging one end and watching for movement elsewhere. (left) Testing a flat cable; these are vulnerable to folding/ kinking, so lay them carefully (right) Using the excellent, but sadly discontinued Vision Test Kit to check AV cables and plugs – can clever PE readers come up with a viable alternative? 16 Practical Electronics | January | 2024 Failing OLEDs Whole batches of radios and Internet radios (for example, from British DAB pioneers Pure and Roberts) were built with OLED displays which are now failing. Without a means to see the settings and tuning options, otherwise perfectly good radios become unusable. This frustration is compounded by the fact that ‘simply’ replacing the display turns out to be tricky, expensive and more trouble than it is worth. However, by chance I discovered that pointing a smartphone camera at the failing OLED display provides a much more readable image on the smartphone screen. With so many old smartphones now languishing in drawers, perhaps someone might rise to the challenge of designing an ageing-OLED viewer for equipment troubleshooting? NEW! 5-year collection 2017-2021 All 60 issues from Jan 2017 to Dec 2021 for just £44.95 PDF files ready for immediate download See page 6 for further details and other great back-issue offers. Ethernet testers are not just for computer/network troublshooting. Ethernet crops everywhere, including home AV systems. Make sure you have one of these handy testers to check cables. Purchase and download at: www.electronpublishing.com tekkiepix pic of the month Mavica – Sony’s pioneering digital camera standard for ‘electronic still video’. Based on the NTSC TV standard. It gave either 25 fully interlaced picture frames, or 50 half-scan pictures of lower resolution. But the technology remained too expensive to rival film, so most companies shelved the idea. Beaten by Fuji Sony’s pioneering digital SLR Mavica offered a range of lenses. S ony obviously liked the name Mavica. In 1974 the company announced the Mavica video recorder, which captured moving pictures on flexible magnetic cards that measured 15 by 20cm, and curved slowly past a scanning video head. Each card could store 10 minutes of colour TV with stereo sound. Sony promised higher density cards with increased recording time, but Mavica was killed by the arrival of Sony’s own Betamax, with several hours recording time from a small cassette. Practical Electronics | January | 2024 Each Mavipak 5cm floppy disc stored up to 50 colour pics. A sony digital first In 1981, Sony dusted off the name and shocked the photographic world by demonstrating a camera, which looked like a conventional SLR (single-lens reflex), but contained an electronic image sensor and miniaturised computer disk recorder. The SLR Mavica recorded analogue TV stills on a 5cm floppy disc coated with pure metal powder and spinning at 3,600 rpm. The pictures replayed through a home TV. In July 1988, 42 electronics and photographic companies agreed a In December 1988, Sony tried to go it alone, launching a consumer version of Mavica in Japan. The full kit cost around £500 and it bombed. By then Fuji had developed a prototype digital camera which used a solid-state memory card instead of disc, which is of course the way modern digital cameras work. However, since memory cards were still expensive, Sony moved on to using a standard 9cm PC floppy to store 40 digital images at very low cost. Practical Electronics is delighted to be able to help promote Barry Fox’s project to preserve the visual history of pre-Internet electronics. Visit www.tekkiepix.com for fascinating stories and a chance to support this unique online collection. 17 We’ve published numerous LC meters that can measure inductance and capacitance, but you might need to know the quality factor (Q) of an inductor, not just its inductance. This Q Meter uses a straightforward circuit to measure the Q factor over a wide range, up to values of about 200. Q Meter T he history of Q Meters goes back to 1934, when Boonton developed the first Q Meter. The Q Meter is a somewhat neglected piece of test equipment these days. Hewlett Packard bought Boonton in 1959 and produced revised versions of their Q Meter. Does anyone still manufacture them? It seems not. You can find a few on the second-hand market; but they fetch prices up to $3000. The HP 4342-A is an excellent unit and is a more modern version of the original Boonton design. My Q Meter design can’t come near the quality or accuracy of HP equipment. It is not designed as a laboratory instrument, but it will give Q measurements up to a value of about 200 with an accuracy of about 10%. Q&A So, what is Q, and why do we need to measure it? It is a measure of the dissipative characteristic of an inductor. High-Q inductors have low dissipation and are used to make Fig.1: a real inductor does not just have pure inductance; it also has parasitic series resistance (Rl) and parallel capacitance (Cp). 18 finely-tuned, narrow-band circuits. Low-Q inductors have higher dissipation, resulting in wideband performance. It can be expressed as: Q = 2π × (Epk / Edis) Where Epk is the peak energy stored in the inductor and Edis is the energy dissipated during each cycle. Let’s consider two passive components, an inductor and a capacitor. The reactance of the inductor is Xl = +jωL. Here, j = √-1, Xl is in ohms and ω = 2πf (f is the frequency). For example, a 10µH coil at 10MHz will have a reactance of +j628W. A capacitor has a reactance of the opposite polarity, ie, Xc = 1/−jωC. To resonate at 10MHz, the capacitor needs a reactance of −j628W, which equates to 25.3pF. By Charles Kosina But inductors and capacitors are not perfect. A practical inductor can be approximated as an ideal inductor with a series resistor. The coil winding will also add a small capacitance across the inductor, as shown in Fig.1. The capacitor is also not perfect but generally has a much smaller inherent resistance, so for this calculation, we can assume it is. The inductor’s Q is defined as Q = Xl/Rl and the -3dB bandwidth of such a tuned circuit is BW = f/Q. So, a tuned circuit with a 10µH coil and a Q of 100 would have a -3dB bandwidth of 100kHz at 10MHz. The Q is important if you’re trying to design something like a bandpass or notch filter. In Fig.2, we have a series-tuned circuit fed by a variable frequency source with frequency f, voltage VS Fig.2: we can calculate an unknown inductor’s Q (quality factor) using this circuit. It is connected in a series-tuned circuit with a capacitance, and that circuit is excited by a sinewave from a signal generator via a known source resistance. Measuring the input and output AC voltages and calculating their ratios allows us to compute the inductor Q, assuming the Q of the capacitance is high. Practical Electronics | January | 2024 and source resistance Rs. At resonance, Xl = −Xc; in effect, a short circuit, so the load on the generator is Rs + Rl. By having a generator with source resistance Rs much lower than Rl, the voltage measured at Vin will be close enough to VS. The current through the circuit will be Is = VS/Rl. Therefore the voltage at the junction of the inductor and capacitor is Vout = Xl × Is. By measuring Vin and Vout, the Q can be calculated as Ql = Vout / Vin. That assumes that the capacitance has been adjusted to achieve peak resonance with the inductance, ie, Xl = −Xc. That can be done by sweeping the capacitance until the peak Vout voltage is reached. The first design challenge is to have an extremely low generator source resistance. If we have a 10µH coil with a Q of 100, at 5MHz, the effective Rl is 3.14W (314W/100). If our source resistance is 0.1W, that will give an error of about 1%. But at 1MHz, Rl becomes 0.628W, and this error blows out to 15%. So using a higher frequency will generally result in a more accurate Q measurement. Low source resistance Boonton solved the source resistance problem by having the generator heat a thermocouple using a wire with a very low resistance, as shown in Fig.3. The voltage generated by this thermocouple was measured by a DC meter which indicated how much current was applied to a 0.02W resistor in series with the external inductor. I have a Meguro MQ-160 Q Meter, essentially a 1968 version of the original Boonton 260-A design, using such a thermocouple and resistor. No transistors in this one; it’s all valves! But for our design, a thermocouple is not practical. The HP design eliminated the thermocouple and instead used a step-down transformer. The Practical Electronics | January | 2024 transformer is fed by a low impedance source, as shown in Fig.4. If our source resistance is 50W, like the output of a typical signal generator, and the turns ratio is 50:1, the effective source resistance is 0.02W (50W/502), exactly what we want. Unfortunately, it is not so simple as it implies a perfect transformer. Losses in the transformer core plus winding resistance conspire against us and push up the source resistance value. We can improve this by feeding the transformer’s primary from the output of an op amp, which has an impedance close to zero. In this case, a turns ratio of 10:1 is adequate as the resultant 100:1 impedance ratio will give an acceptable load to the op amp. This is what I have used in my design. The transformer is a ferrite toroid of 12mm outside diameter. The primary is 10 turns of enamelled wire, while the ‘one turn’ secondary is a 12mm-long tapped brass spacer through the centre of the toroid. The effective RF resistance of this spacer is extremely low, and the source resistance is then mainly a function of the ferrite material and the primary winding resistance. Table 1 – frequency versus signal source impedance/spacer Frequency Brass Steel 0.1-1MHz ~0.00W 0.02W 2MHz not tested 0.016W 5MHz 0.03W 0.13W 10MHz 0.07W 0.20W 15MHz 0.09W not tested 20MHz 0.15W 0.22W 25MHz 0.10W 0.17W Circuit description The full circuit of my Q Meter is shown in Fig.5. We require a signal generator with an output of about 0dBm (1mW into 50W or 225mV RMS). You can use just about any RF signal generator. There didn’t seem to be much point in building the generator into the Q Meter since, if you’re building a Q Meter, you likely already have an RF signal generator. I’m using my AM/FM DDS Signal Generator that was described in the May 2023 issue of PE. The generator feeds a sinewave into CON1, which is boosted by op amp IC2a. This is a critical item in the design, as it needs to have a high gain bandwidth (GBW) and slew rate, as well as the capability to drive a low impedance. The Texas Instruments OPA2677 has a GBW of 200MHz, a slew rate of 1800V/µs and can drive a 25W load, which gives us enough output voltage swing up to 25MHz. The toroidal transformer core is a critical part of the design. I tested a Fair-rite 5943000301 core which is readily available from several suppliers. I wound it with 10 turns of 0.3mm-diameter enamelled copper wire. A heavier gauge (up to about 0.4mm) may be slightly better, but there has to be enough room in the centre for the spacer to pass through. I then calculated the source impedance by measuring the no-load output voltage followed by a 1W load. I did this for several frequencies, and the results are shown in Table 1. Below 1MHz, there was no measurable difference between no load and a 1W load, so the source impedance must be well below 0.01W. Core losses likely account for the higher source resistance as frequency increases, but the results are quite adequate. Brass spacers are recommended (and will be supplied in kits) due to their superior performance here, at least for the one through the toroid. The DC output of op amp IC2a is zero or very close to zero, so why do Fig.3: one method of measuring Q involves current sensing via monitoring the temperature of a resistance wire. It has the advantage of keeping the source impedance low, and no complicated shuntsensing circuitry is required. Fig.4: we need an RF signal source with an extremely low (but known) source resistance for our Q Meter. Since that is difficult to achieve by itself, feeding the signal through a low-loss stepdown transformer greatly reduces the actual source impedance, as seen by the load. 19 Digital Q Meter Fig.5: eight relays switch capacitors in parallel to vary the resonant circuit capacitance from around 40pF (the stray capacitance) to 295pF. The signal from the RF generator is amplified by op amp IC2a and fed through step-down transformer T1 to the resonant circuit. The input signal level is monitored via precision rectifier IC2b while the output signal is rectified using D3 and amplified by IC3a. we need a 10µF capacitor in series with the transformer? Since the DC resistance of the primary is a fraction of an ohm, the slightest offset 20 voltage in the op amp output could send a high direct current through the toroidal transformer primary and overload the output. IN this design, that possibility is eliminated with AC coupling. The tuning capacitor is another essential part. My Meguro MQ-160 Q Practical Electronics | January | 2024 Meter has a 22-480pF variable capacitor, typical of the tuning capacitors used in valve radios. They are available on sites like eBay, but they do Practical Electronics | January | 2024 tend to be rather large and can be surprisingly expensive. The only easy-to-get variable capacitor is the sort with a plastic dielectric for AM radios. But once you get above the broadcast band, they are very lossy, with a poor Q, and entirely unsuitable. So instead, I designed a ‘digital capacitor’ with eight relays switching in capacitors with values in a binary sequence of 1, 2, 4, …..128pF. As these are not standard values, some are made up of two capacitors in parallel. For example, 32pF is 22pF in parallel plus 10pF. Combining these allows the capacitance to be adjusted in 1pF steps from 0pF to 255pF. The measured stray capacitance due to the tracks, relays and so on amounts to 40pF, so the tuning range is 40-295pF. My LC meter shows that it tracks reasonably accurately. All capacitors are not created equal, so I have used somewhat expensive high-Q RF capacitors, available from element14, Mouser, Digi-Key and other good suppliers. Not all these capacitors have a close tolerance; some are ±2%, which detracts from the accuracy. So it isn’t a ‘real’ variable capacitor but it has the advantage of not needing a calibrated dial and a slow-motion vernier adjustment. Rather than measuring the very low voltage on the secondary side of the transformer, it is more practical to measure the primary side, and for the Q calculation, divide this by 10. I verified this assumption by checking that the voltage ratio corresponded to the turns ratio within measurement accuracy from 100kHz to 25MHz. A precision half-wave rectifier is formed using op amp IC2b in the classic configuration. By placing the rectifier diodes in the negative feedback network of the op amp, their forward voltages are effectively divided by the (very high) open-loop gain of the op amp. On positive excursions of the output pin of IC2b, the 330nF capacitor at TP3 is charged up through diode D1. The extra diode (D2) is needed becuae without it, negative excursions would saturate the op amp and lead to slow recovery, limiting its frequency range. Both diodes are 1N5711 types for fast switching. The output of IC2b is amplified by IC3b, and the resulting filtered DC voltage at TP4 is about 1.9V. The secondary voltage of the transformer is typically 200mV peak-topeak or about 70mV RMS. With a Q of 100, the voltage output at the junction of the inductor and tuning capacitor would be 20V peak-to-peak or 7V RMS. 21 Fig.6: the PCB uses mostly SMD components for compactness, although none are particularly small. The orientations of the following components are important: all relays, ICs and diodes, plus the Arduino Nano. ZD1, IC4, CON3 and associated parts form the optional debugging interface. That is not a suitable voltage to apply to the input of an op amp! So I used schottky diode D3 as a half-wave rectifier feeding a high-­ impedance (10MW/1.5MW) voltage divider. The voltage drop in the diode only introduces a small error in the measurement. The voltage at the junction of this divider is buffered and amplified by IC3a, a TSV912 op amp with an extremely high input impedance – the input bias current is typically 1pA. Switch S1 changes the gain of this op amp for the low and high Q ranges, with the low range giving a gain of 8.3 for Q values of up to 100. 22 On the high range, the gain of this stage drops to 1.7. Power supply and control A MAX660 switched capacitor voltage inverter (IC1) provides a nominally −5V supply to the OPA2677 (IC2). This is needed for proper operation of the half-wave precision rectifier ( IC2b) since the voltage at its input can swing below ground. The MAX660 is not a perfect voltage inverter, and with the current drain of the OPA2677, its output is about −3.6V, but that is adequate. The rest of the circuit operates from a regulated +5V DC fed in externally – for example, using a USB supply. An Arduino Nano module is used as the controller. This is a readily-­ available part from many suppliers at a reasonable price. Two analogue inputs are used for measuring the voltages, eight digital outputs switch relays, the two I2C serial lines drive the OLED, and there are inputs for the control rotary encoder and LOW/ HIGH switch sensing. The rotary encoder (EN1) is used to adjust the ‘digital capacitor’ value; its integral pushbutton switch toggles between steps of 1pF and 10pF. As usual with my designs, I have added a simplified RS-232 interface using hex schmitt-trigger inverter IC4 to aid code debugging. IC4, ZD1 and the two associated resistors can be left out unless you want to use the debugging interface. Eight 2N7002 N-channel MOSFETs (Q1-Q8) drive the relay coils, while eight diodes across the relay coils (D6-D13) suppress any switching transients. The resonant frequency tuning is done by selecting an appropriate frequency from the external signal generator and adjusting the variable capacitance value. Ideally, the peaking should be done with an analogue meter, but I have provided an onboard LED (LED1), the brightness of which depends on the Vout voltage. It’s a simple enough procedure to adjust the capacitance to achieve maximum brightness. The third line of the OLED also shows the output voltage of IC3a, which can be used to accurately achieve resonance too. Connector CON5 drives an optional external 0-5V moving-coil meter. You can add such a meter if a larger-­than-specified enclosure is used to house the PCB. The power supply is a standard 5V USB charger. I have not included reverse polarity protection, but an off-board 1A schottky diode (eg, 1N5819) could be added in series if desired. Construction The construction uses two PCBs (see Figs.6 and 7). The main one has all the electronics while the other has the screw terminals for the DUT and external capacitor. It is also used as a front panel and has a rectangular cutout for the OLED, holes for the controls and lettering. It is designed to fit in a RITEC 125 × 85 × 55mm enclosure (for example, one sold by Altronics as H0324, but plenty of other vendors will have similar boxes). Practical Electronics | January | 2024 The top board/front panel is 98 × 76mm and fits snugly into the recess in the clear lid of the enclosure. This board could be used as a template for accurately drilling the holes in the clear lid. But other enclosures may be used as long as they have the same or slightly greater dimensions as the H0324. For those wishing to add the 0-5V moving-coil meter, this requires an additional width of 45mm. A suitable 158 × 90 × 60mm enclosure is available from AliExpress suppliers at a reasonable price, but do remeber that delivery can take quite a few weeks. Most components on the PCB are surface-mount types, but there are no fine-pitch ones, which simplifies construction. Solder the four SOIC chips first, then all the passives, which are mostly M2012/0805-size devices (2.0 × 1.2mm). The relays take a bit of care to ensure they are square on the board so that it looks neat. On the opposite side of the board are eight 1N4148 equivalent diodes; ensure they are installed with the correct polarity, with the cathode stripes to the side marked ‘K’. After the SMDs, add the throughhole diodes, which have a 7.6mm (0.3-inch) pitch, then the rotary encoder, switch and LED. Use a 5mm plastic spacer for the LED, so it is flush with the back of the front panel. Wind ten turns of the specified enamelled copper wire onto the toroidal core, taking care that the turns are equally spaced around the circumference, to the extent possible, and the ends line up with the two pads marked PRIM on the PCB. Carefully attach the toroid so that it is centred on the mounting hole. Attaching the spacer to the board makes that easier. It may be anchored in place by an insulated wire across the two pads on the opposite side. It is not a shorted turn because only one side of this wire is connected to the ground plane. I recommend fitting socket strips for mounting the Arduino Nano module as they make replacing a faulty module easy (I have blown up a couple in the past!). The OLED screen also plugs into a 4-pin socket strip and is held in place by two 15mm-long M2 or M2.5 screws through 8mm untapped spacers. Carefully slide off the plastic strip on the four pins of the OLED so that it sits lower. The board must be thoroughly cleaned with board cleaner. There are Practical Electronics | January | 2024 high impedances throughout the circuit, so be aware that leakage through flux residue would affect its operation – you must remove that residue. Testing Once the board has been fully assembled, cleaned and inspected, but before it is mounted in the case, attach the four 12mm spacers – but not the front panel board – and connect the 5V supply. The OLED should show an initial message with the firmware version number. Using a coax cable, feed in a sinewave from a signal generator at about 1MHz. An oscilloscope probe on TP1 should show a clean sinewave, with an output of about 2V peak-to-peak. If the output of the signal generator is too high, you will get flattening on the negative half cycle. In that case, back off the level for a clean sinewave. Transfer the ‘scope probe to the top of the spacer that passes through the toroid, and the voltage should be one-tenth of that measured at TP1. Measure TP4 using a DC voltmeter; Only the Arduino Nano, headers and eight diodes are on the underside of the Q Meter PCB. Note how the windings for T1 are spaced evenly around it. 23