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AUDIO
OUT
AUDIO OUT
L
R
By Jake Rothman
Back to the buffers – Part 4
T
his month, we’ll wrap up the
series on discrete op amp/buffer
designs. The complexity of a full
discrete op amp is not really required for
a buffer and the simple three-transistor
circuits given (PE, April 2024, Fig.24,
Fig.27) are usually sufficient. However,
sometimes op amp characteristics, such
as a low output offset are occasionally
needed. The high open-loop gain of a
full op amp (discrete or monolithic) also
enables a very low output impedance to
be obtained. There are some expensive
dedicated buffer chips available, such
as the LM302, which employs 18
transistors. So, I thought a ‘halfwayhouse’ discrete solution based on an
op amp input stage with the voltage
amplifier stage (VAS) omitted would be
worthy of some R&D.
Op amp buffer design
My resulting design is a two-stage op
amp, sometimes called a ‘Schlotzaur
circuit’. It has an open-loop gain of
around 6000 if constant current loads
are used. To start, the discrete op amp
board from the October 2023 article was
used with a long-tailed pair (LTP) stage
followed by a class-A emitter follower.
A possible problem with this topology
using 100% negative feedback, is that
the operating level of the input stage
is the same as the output. Indeed, this
was a problem, with the distortion level
increasing in proportion to the output
voltage, as shown later.
Development trials
and tribulations
Originally, the output stage of the first
discrete op amp which I used as a buffer
just consisted of an emitter follower
(TR3) and current sink running at 10mA
(TR7) – see Fig.35. This was adequate
for high impedance loads, but not for
full output into 600Ω. The solution
was a three-transistor follower and a
modulated current source. The first
circuit tried was to simply connect the
three-transistor buffer to the op amp
input stage. A successful circuit design
58
strategy is to start by combining circuit
blocks you already know work. If it
doesn’t work, it’s easier to find the fault.
There was a hard offset upon turn-on
because I forgot that the omitted voltageamplifier stage was inverting. To get
round this, the negative feedback had to
be applied to the non-inverting terminal
OA+ on the input stage. Guess what? It
still didn’t work. Then I remembered the
current mirror had to be turned round
as well. One side of the mirror is output
and the other, the diode-connected
transistor side (TR5 on the original op
amp circuit (PE, October 2023, Fig.15)),
is for current sensing. Having done this
necessary ‘flipping’, the circuit worked.
Next, I needed to do some optimisation.
Oscillation
subtle low-frequency rise in THD that
occurs with capacitors, for example.
To fix the oscillation, I placed a
capacitor (CComp) across the collectors
of the long-tailed pair. I used 470pF out
of the ‘mixed cap’ drawer which fixed
the oscillation problem. I then checked
for slewing distortion which occurred
at a too low a frequency of 40kHz, so
I then reduced the capacitor to 150pF
– but then some other unwanted highfrequency signal appeared. Fortunately,
this was not an oscillation, but 470kHz
RF noise emitted from a new Metcal
soldering iron I had recently started
using. Unfortunately, this new problem
also had me going round in circles for
a couple of days after sudden rises in
distortion on various amplifiers I was
testing. The problem was solved by
moving the soldering iron away from
the test circuit, at which point I was
relieved to determine that the 150pF
As is usual with new circuits with
negative feedback, it oscillated when
first turned on. It was barely visible
on the scope as a
thicker trace, but
+25V
very apparent on the
19mA
R3
R4
R7
distortion analyser,
PR1
200Ω
200Ω
10kΩ
5kΩ
with a reading of
0.02% THD+N
4.8mA
a t 1 V rms ( t o t a l
DC offset
harmonic distortion
TR5
TR4
BC556
BC556
TR3
plus noise). I knew
BC546
from before that
*Compensation
R22
both circuit blocks
capacitor
47Ω
CComp*
were capable of
VO
VIN
150pF
at least ten-times
TR2
TR1
BC546 BC546
better distortion.
100%
R18
22kΩ
Negative
R1
R2
One reason I pursue
feedback
110Ω
110Ω
vanishingly low
0V
2mA
2mA
distortion figures
TR7
BC546
4mA
in my circuits,
R6
2.2kΩ
even though the
TR6
BC546
improvements may
10mA
be inaudible, is to
LED1
R8
1.7V
R5
Red
clear the view for
100Ω
240Ω
other problems and
–25V
non-linearities that
19mA
can hide behind
a h i g h g e n e r a l Fig.35. Initial discrete op-amp-style buffer. This is reconfigured
THD floor. These to compensate for the omission of the inverting VAS stage.
can be low-level Note component numbering refer to the discrete op amp PCB
oscillations or the (PE, November, 2023).
Practical Electronics | May | 2024
Fig.36. The effect of using an output MOSFET for the JFET discrete op-amp-based
buffer circuit was to reduce the distortion compared to a Darlington (TR3) in Fig.37.
R2
200Ω
PR1
5kΩ
2mA
DC offset
2mA
5mA
TR4
*Compensation
capacitor
VIN
R7
10kΩ
19mA
R8
180Ω
1.78V
TR5
TR4/5
HN1A01F
TR1/2
2SK2145
R3
200Ω
+25V
D
C1*
150pF
TR1
R1
1MΩ
+
D
S
R4
240Ω
R10
47Ω
VO
100%
Negative
feedback
4mA
TR3
BC546
TR6
BS170 D
TR2
S
0V
C3
10µF
50V
R5
2.2kΩ
C2
100nF
LED1
Red
R6
2.2kΩ
TR7
BC546
10mA
1.7V
R9
100Ω
19mA
–25V
Fig.37. Final discrete MOSFET op-amp-based buffer circuit with JFET input and MOSFET
output. (Note: components renumbered, coupling/decoupling capacitors omitted.)
Fig.38. Building the MOSFET op-ampbased buffer on the discrete op amp
PCB (PE, November, 2023). This is how
prototypes can look. Note that individual
devices can be plugged in to sockets for
changing devices during testing.
capacitor was sufficient to fix the
oscillation problem.
Another cause of ‘raised distortion’
was a bench power supply placed too
close to the Audio Precision audio
analyser. I found magnetic coupling
between the power supply’s large,
laminated transformer and the audio
analyser’s output transformers. About
200mm clearance was required.
Since the output stage was enclosed
in a negative feedback loop, I found the
value of the sense resistor (R8, Fig.37)
was much less critical for minimum
distortion than in the three transistor
circuits. I then set it for maximum output
swing into a 600Ω load.
Next, I decided to run the input stage
at double the current (2mA per each
transistor) to increase the slewing, give
greater drive to the output stage and to
get more transconductance out of JFETs
if these are used for the long-tailed pair.
Component selection
Following on from my experience with
parts for the discrete op amp, I strongly
suspcted dual transistors would give
the best results – and they did. Using
JFETs for the long-tailed pair revealed
them to be more sensitive to loading
from the output stage. I tried an MPSA29
Darlington transistor for the emitter
follower (TR3) and the distortion
dropped by 75% compared to a BC546B.
Fig.39. Increasing the output level for
the FET (JFET input, MOSFET output)
op amp buffer in Fig.37 in 1Vrms steps
to 12Vrms into a 600Ω load increases
the distortion as expected. Note the
rapid rise of distortion at 20Hz to 1.3%
at 12V due to the 10µF 50V tantalum
bead modulation capacitor (C3) being
stressed. Changing this device to
a 22µF 50V metal-cased tantalum
reduced it to 0.025%.
Practical Electronics | May | 2024
59
However, some sort of curvature
cancellation magic happened when I
put in a BS170 MOSFET with currentsink modulation. Using this device the
circuit achieved the excellent figure of
0.001% THD+N, as shown in Fig.36.
My final circuit is shown in Fig.37, and
the messy prototype construction of the
discrete op amp PCB in Fig.38.
Distortion depression
Fig.40. Using the original full discrete op amp as a buffer suffers from rising highfrequency distortion. Note that the top curve is for the output just before clipping at
17Vrms.
Fig.41. Distortion for three-transistor buffer just before clipping (Fig.24). These curves
are taken in steps from 1V to 15Vrms into 600Ω. Clipping is just beginning to occur at
15Vrms, hence the sudden jump in distortion.
Fig.42. Distortion plot for JFET single-ended buffer in Fig.27 at 1Vrms, 4Vrms and
8Vrms respectively.
60
However, this low distortion was only for
an output signal of 1Vrms (2.8Vpk-pk) into
600Ω. I needed to check the distortion
at different output levels all the way up
to clipping. I increased the output level
in 1Vrms steps on the Fig.37 FET op
amp buffer giving the curves shown in
Fig.39. What looked good at 1Vrms didn’t
look so good at higher test levels up to
clipping. There was a rise in distortion
at the low-frequency end at 12Vrms due
to the modulation capacitor C8 getting
stressed.
I then tested the original full discrete
op amp (PE, October 2023, Fig.15) wired
as a buffer with 100% negative feedback
and this gave a large high-frequency
(10kHz) distortion hump of 0.026% at
16Vrms, shown in Fig.40. This is because
a lot of compensation capacitance is
required to make this topology unitygain stable. This causes a problem with
slewing when driving the capacitance
and there is little negative feedback at
high frequencies left for output stage
distortion reduction. It did give a large
output of 17Vrms (48Vpk-pk) just beginning
to clip with ±25V rails, 2Vrms more than
the single-ended three transistor circuit
for the same rail voltage.
Back to the threetransistor buffer
I then checked the three-transistor buffer
circuit shown in Fig.24 which gave better
performance than the op amp version at
the higher test levels (above 1Vrms), as
shown in Fig.41.
I can’t leave things alone and
decided to replace R7 with a more
stable constant-current source using
a 1mA CRD (current-regulator diode).
The distortion dropped by a third to
0.0015% at 8Vrms and the low-frequency
distortion increase vanished. At 1Vrms
the distortion was flat over the whole
band at 0.0005%, a useful upgrade.
Sadly, this was not so effective on the
JFET version which also needed the
sense resistor R8 adjusting to 470Ω. I
was hoping this constant-current mod
would reduce the distortion variation
between different individual JFETs. It
didn’t. If I was a total obsessive, I would
make R8 a preset adjustment tweaked for
each individual JFET. Fig.42 shows the
distortion at different levels produced
Practical Electronics | May | 2024
Fig.43. Spectrum of distortion harmonics
produced by the single-ended JFET
buffer in Fig.27 driven at 1kHz 0dB. Note
the relatively large amount of second
harmonic (2kHz) which is 90dB down
relative to 1kHz. The third harmonic is
122dB down. Since it is a simple circuit
with low feedback the higher harmonics
are invisible, buried in noise.
by the JFET single-ended three-transistor
buffer. However, the bulk of distortion
generated is second harmonic, as shown
in the spectrum analysis plot in Fig.43
produced using fast Fourier transform
analysis. FFT is a useful feature of the
Audio Precision analyser, TiePie scopes
and various software packages. This type
of distortion at this level is inaudible, but
at high levels can even sound nice, so
the variation between JFETs is nothing
to worry about.
C5
100nF
It’s for real
4.8mA
R3
10kΩ
VIN
C7
100nF
C1
470nF
R2
1kΩ
R1
100kΩ
C2
100pF
R4
130Ω
Ib
Ib
TR2
BC456B
TR5
ZRX651
R8
100Ω
4.5mA
TR1
BC556B
Iq
4.2mA
R6*
6.8Ω
al
rm
he
T
link
*Set Iq
0V
Th
erm
al
link
R10
22Ω
+
C3
22µF
10V
R11
22Ω
92mV
C4
47µF
Bipolar
R12
47Ω
VO
R7*
6.8Ω
R9
100Ω
0V
D3
1N4148
C6
100nF
D4
1N4148
TR4
BC456B
R5
130Ω
TR6
ZTX751
4.2mA
13.5mA
–25V
Fig.45. ‘Diamond’ buffer. An unusual circuit with two parallel, but opposite polarity input
emitter follower stages. These then drive a complementary class-AB emitter-follower output.
Fig.44. ‘Cupboard based programming language’ for analogue
circuits. This shelf of passive components comprises 650
Farnell plastic drawers alone. (These cost more than the parts
in some cases!)
Practical Electronics | May | 2024
D2
1N4148
TR3
BC556B
+25V
+
It is possible to use simulation for this
type of R&D and it is great for checking
whether a circuit will basically work,
avoiding the latch-up scenario I had
earlier. However, I dislike simulation
for distortion analysis because it tends
to underestimate problems. Also, I’ve
been doing practical circuits since I
was 12 and I’m now 61, so I stick to
what I’m good at! It takes me longer to
input the schematic into LTSpice than
it does to build it. I can pull out any
component from my vast ‘CupboardBased Programming Language’
(CBPL, shown in Fig.44) quicker than
downloading any component model
on-line. When I need a simulation I can
always get a friend with the necessary
10,000-hours practice time to do a
skills swap with me. The catch with
my CBPL method is that the file size
is 1500 square feet. You need to have
bought a house before 1995 or have had
a big inheritance to install it.
0V
D1
1N4148
13.5mA
Fig.46. This Avondale Audio active crossover board uses four
diamond buffers with a similar proprietary circuit to Fig.45. The
transistors are thermally coupled to ensure a stable quiescent
current. Note there are lots of SMT components under the board.
61
Diamond buffer
C9
C5
LED1
C7
+VE
R5
R6
R12
GND
-VE
C4
R7
C6
R10
ZD1
C3
R3
I/P
GND
SGL SUPPLY
LINK
The Diamond buffer is an interesting,
if poorly named circuit, which I first
came across in an Elektor oscillator
design (February 1979). It is a classAB push-pull mirror-image circuit
that has curvature cancellation, giving
very low distortion of 0.0018% at 1Vrms
into 600Ω with no overall feedback.
The transistor input bias currents also
cancel out, assuming good Hfe matching.
Despite its name I can’t see a ‘diamond
shape’ in it; rather it has a ‘crossover’
like the classic multi-vibrator. I don’t
know what the ‘diamond’ refers to, its
name seems to derive from Burr Brown
blingy marketing. The circuit in Fig.45
needs six transistors, which is rather
a lot for a buffer. The Elektor version
was simplified down to four transistors
by using bootstrapping for the input
transistors’ collector loads. Note the
thermal linking of the transistors to
reduce drift. A practical application
is for the buffers in active filters, such
as the active speaker crossover shown
in Fig.46.
C1
R8
TR1
R9
O/P
GND
R1
R4
TR2
R11
TR3
C10
R2
C8
R13
C2
R14
Fig.47 Held over from last month, the overlay diagram for the PCB for last month’s
three-transistor discrete buffer circuit.
Software sorrow… and a little serendipity
During the test phase for the discrete buffer I thought I had a fault with my
AP audio analyser – the output drive level seemed to be fixed at 1Vrms, but
I was wrong. Buried in the manual (six A4 ring binders), I found – after
a year of use – that the analyser was running a fixed test routine (macro)
each time I clicked on the little THD icon. If I went to the sweep menu
and clicked ‘start with append’ I could then change the output drive level
with at least three more entries and clicks. I was amazed to find the unit
could actually deliver 26Vrms (36.9Vpk-pk) at 0.0002% THD into a 600Ω
load. This is very useful for performing component stress tests. I was even
able to produce distortion in resistors! But that’s for another column. I’m
sure there are many other settings I could use better, does anyone out there
know how to use an AP SYS-2712 to its full potential?
PCB
A PCB for the discrete buffer is available
from the PE PCB Service – layout, Fig.47.
STEWART OF READING
Fluke/Philips PM3092 Oscilloscope
2+2 Channel 200MHz Delay TB,
Autoset etc – £250
LAMBDA GENESYS
LAMBDA GENESYS
IFR 2025
IFR 2948B
IFR 6843
R&S APN62
Agilent 8712ET
HP8903A/B
HP8757D
HP3325A
HP3561A
HP6032A
HP6622A
HP6624A
HP6632B
HP6644A
HP6654A
HP8341A
HP83630A
HP83624A
HP8484A
HP8560E
HP8563A
HP8566B
HP8662A
Marconi 2022E
Marconi 2024
Marconi 2030
Marconi 2023A
HP 54600B Oscilloscope
Analogue/Digital Dual Trace 100MHz
Only £75, with accessories £125
(ALL PRICES PLUS CARRIAGE & VAT)
Please check availability before ordering or calling in
PSU GEN100-15 100V 15A Boxed As New
£400
PSU GEN50-30 50V 30A
£400
Signal Generator 9kHz – 2.51GHz Opt 04/11
£900
Communication Service Monitor Opts 03/25 Avionics
POA
Microwave Systems Analyser 10MHz – 20GHz
POA
Syn Function Generator 1Hz – 260kHz
£295
RF Network Analyser 300kHz – 1300MHz
POA
Audio Analyser
£750 – £950
Scaler Network Analyser
POA
Synthesised Function Generator
£195
Dynamic Signal Analyser
£650
PSU 0-60V 0-50A 1000W
£750
PSU 0-20V 4A Twice or 0-50V 2A Twice
£350
PSU 4 Outputs
£400
PSU 0-20V 0-5A
£195
PSU 0-60V 3.5A
£400
PSU 0-60V 0-9A
£500
Synthesised Sweep Generator 10MHz – 20GHz
£2,000
Synthesised Sweeper 10MHz – 26.5 GHz
POA
Synthesised Sweeper 2 – 20GHz
POA
Power Sensor 0.01-18GHz 3nW-10µW
£75
Spectrum Analyser Synthesised 30Hz – 2.9GHz
£1,750
Spectrum Analyser Synthesised 9kHz – 22GHz
£2,250
Spectrum Analsyer 100Hz – 22GHz
£1,200
RF Generator 10kHz – 1280MHz
£750
Synthesised AM/FM Signal Generator 10kHz – 1.01GHz
£325
Synthesised Signal Generator 9kHz – 2.4GHz
£800
Synthesised Signal Generator 10kHz – 1.35GHz
£750
Signal Generator 9kHz – 1.2GHz
£700
HP/Agilent HP 34401A Digital
Multimeter 6½ Digit £325 – £375
62
17A King Street, Mortimer, near Reading, RG7 3RS
Telephone: 0118 933 1111 Fax: 0118 933 2375
USED ELECTRONIC TEST EQUIPMENT
Check website www.stewart-of-reading.co.uk
HP33120A
HP53131A
HP53131A
Audio Precision
Datron 4708
Druck DPI 515
Datron 1081
ENI 325LA
Keithley 228
Time 9818
Marconi 2305
Marconi 2440
Marconi 2945/A/B
Marconi 2955
Marconi 2955A
Marconi 2955B
Marconi 6200
Marconi 6200A
Marconi 6200B
Marconi 6960B
Tektronix TDS3052B
Tektronix TDS3032
Tektronix TDS3012
Tektronix 2430A
Tektronix 2465B
Farnell AP60/50
Farnell XA35/2T
Farnell AP100-90
Farnell LF1
Racal 1991
Racal 2101
Racal 9300
Racal 9300B
Solartron 7150/PLUS
Solatron 1253
Solartron SI 1255
Tasakago TM035-2
Thurlby PL320QMD
Thurlby TG210
Modulation Meter
£250
Counter 20GHz
£295
Communications Test Set Various Options
POA
Radio Communications Test Set
£595
Radio Communications Test Set
£725
Radio Communications Test Set
£800
Microwave Test Set
£1,500
Microwave Test Set 10MHz – 20GHz
£1,950
Microwave Test Set
£2,300
Power Meter with 6910 sensor
£295
Oscilloscope 500MHz 2.5GS/s
£1,250
Oscilloscope 300MHz 2.5GS/s
£995
Oscilloscope 2 Channel 100MHz 1.25GS/s
£450
Oscilloscope Dual Trace 150MHz 100MS/s
£350
Oscilloscope 4 Channel 400MHz
£600
PSU 0-60V 0-50A 1kW Switch Mode
£300
PSU 0-35V 0-2A Twice Digital
£75
Power Supply 100V 90A
£900
Sine/Sq Oscillator 10Hz – 1MHz
£45
Counter/Timer 160MHz 9 Digit
£150
Counter 20GHz LED
£295
True RMS Millivoltmeter 5Hz – 20MHz etc
£45
As 9300
£75
6½ Digit DMM True RMS IEEE
£65/£75
Gain Phase Analyser 1mHz – 20kHz
£600
HF Frequency Response Analyser
POA
PSU 0-35V 0-2A 2 Meters
£30
PSU 0-30V 0-2A Twice
£160 – £200
Function Generator 0.002-2MHz TTL etc Kenwood Badged
£65
Function Generator 100 microHz – 15MHz
Universal Counter 3GHz Boxed unused
Universal Counter 225MHz
SYS2712 Audio Analyser – in original box
Autocal Multifunction Standard
Pressure Calibrator/Controller
Autocal Standards Multimeter
RF Power Amplifier 250kHz – 150MHz 25W 50dB
Voltage/Current Source
DC Current & Voltage Calibrator
£350
£600
£350
POA
POA
£400
POA
POA
POA
POA
Marconi 2955B Radio
Communications Test Set – £800
Practical Electronics | May | 2024
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