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Precision Electronics part three
Precision
Electronics
Part 3: difference & instrumentation amplifiers
In this third article in this series, we will further develop our precision current
measuring circuit. We will consider how to sense the current if the shunt was in the
positive line instead of referenced to circuit ground.
By Andrew Levido
Y
ou will recall that previously, we
sensed a 0 to 1A current using a
100mW resistor, one side of which
was connected to circuit ground. We
amplified the resulting voltage by a
factor of around 25 to get a ground-
referenced signal of about 2.5V fullscale, which we could apply to an
analog-to-digital converter (ADC) to
make the measurement.
We are assuming there is a microcontroller in the circuit that can trim
out much of the fixed offset and gain
error, leaving us with a trimmed precision of around ±0.04% at 25°C with
about ±0.075% additional error over
the 0–50°C operating temperature
range. We deemed this overall precision of just over 0.1% ‘good enough’
for our purposes.
In practice, we often want to sense
the current in the positive leg of the
circuit, as shown in Fig.1. The reason
is that it is possible (sometimes even
unavoidable) that the grounds of both
the source and the load are connected to a common potential, such as
mains Earth.
If this were to happen with the original circuit, the sense resistor would
be shorted out, so measuring the current would be impossible.
Moving the shunt resistor to the
positive line solves that problem but
introduces another. One terminal of
In cases like this where we have two
sense terminals, we refer to the voltage between them as the differentialmode voltage (Vdm) and the voltage at
the terminals with respect to ground as
the common-mode voltage (Vcm). This
is shown diagrammatically in Fig.2.
The differential voltage of interest
(Vdm) is ‘riding on’ the common-mode
voltage (Vcm) that we want to ignore.
For ground-referenced signals, the
common-mode voltage is zero (in an
ideal world, anyway).
The output of the generalised conditioning circuit block will be a voltage that is the sum of the differential-
mode input (V dm ) multiplied by a
differential-m ode gain (G dm), along
Fig.1: to measure current with a
sense resistor in the positive line,
we need to extract the relatively
small differential signal from the
larger common-mode signal.
Fig.2: to achieve what we need in
Fig.1, the “Signal Conditioning” box
needs to amplify Vdm with a high gain
(Gdm) but minimise the contribution of
Vcm, meaning Gcm should be kept low.
26
the resistor is sitting at the load’s
positive supply voltage (up to 20V in
our example), while the other is up
to 100mV higher, depending on the
current through it. We are interested
in amplifying only the difference in
voltage between these two points, not
the much larger voltage on which it
is floating.
We also want the resulting amplified signal to be referenced to circuit
ground so it’s within the ADC’s range,
and so we don’t need to use a differential ADC to measure it.
Differential and common
mode signals
with the common-mode input (Vcm)
multiplied by the common-mode gain
(Gcm). We usually want Gcm to be zero
(or as close to it as we can practically
get) so that the unwanted common-
mode voltage is rejected.
We describe the degree to which a
circuit like this rejects common mode
signals as the common-mode rejection ratio (CMRR). This is the ratio
of the differential-mode gain to the
common-mode gain (Gdm ÷ Gcm) and
is usually expressed in decibels, calculated as 20log10(Gdm ÷ Gcm).
You have probably guessed by now
that any hope of perfectly rejecting
common-mode signals (ie, achieving
an infinite CMRR) is just a pipe dream.
The harsh reality of electronics design
means we always have to put up with
something less than perfection.
Difference amplifiers
One of the most common ways to
amplify a small differential signal
riding on a large common-mode voltage is to use a difference amplifier like
that shown in Fig.3.
Two pairs of matched resistors (R1a
= R1b & R2a = R2b) and an op amp
form an amplifier with some very interesting characteristics.
This general form of difference amplifier (with separate sense and reference terminals) is a very flexible
Fig.3: the classic difference amplifier
using an op amp and four resistors
is a very useful and flexible circuit.
Usually, the value of R1a is the same
as R1b and R2a the same as R2b.
Practical Electronics | March | 2025
Difference & instrumentation amps
circuit that can be used to implement
a wide variety of functions, as shown
in Fig.4.
All those circuits use a difference
amplifier with unity gain (R1a = R1b
= R2a = R2b).
The terms “difference amplifier”
and “differential amplifier” are often
used interchangeably. I am using the
former term to describe any amplifier
in which the output is proportional
to the difference between the input
voltages.
Some sources use “differential amplifier” as the general term and “difference amplifier” to describe the specific
configuration where the differential-
mode gain is equal to one (eg, pemag.
au/link/ac1h).
To add to the confusion, the terms
“differential amplifier” and “fully differential amplifier” are both used to
describe op amps with complimentary
positive and negative outputs. These
are specialised devices are normally
used to drive high-speed twisted pairs
or differential input ADCs
Getting back to the device itself, connecting the sense terminal of a difference amplifier to the output and the
reference terminal to ground produces
the familiar configuration illustrated
in Fig.5. This has a differential-mode
gain of Gdm = R2 ÷ R1 (where R1 = R1a
= R1b and R2 = R2a = R2b).
The common-mode gain of the difference amplifier would be zero if the
op amp was ideal and the resistor
matching was perfect. If you build a
difference amp with a typical op amp
with an open loop gain of 100,000 and
1% resistors, the CMRR would be in
the region of 34dB.
This means you would see about
1 /50th of the common-mode voltage
at the output. That would equate to
400mV in our example, almost half
of the differential-mode signal we are
interested in!
We can do better with matched resistors. For example, using the ACASA
range of resistor arrays we used last
time (matched to within 0.05%), we
would have a CMRR in the order of
60dB. That still means we would
see a common-mode voltage of up to
20mV at the output, which is clearly
not good enough for our application.
Yo u c a n b u y i n t e g r a t e d d i f ference amplifiers with on-board
laser-t rimmed resistors that have
CMRR values in the 80-100dB range
at modest cost. If we used one of
these, say with a CMRR of 90dB, the
common-mode voltage at the output
would be just 632µV. That is pretty
good, but it still represents a 0.025%
error, which will have to be added to
the other errors.
There is a bigger problem, however.
Off-the-shelf difference amplifiers are
typically only available with gains up
to about 10, with most having a gain
of just one or two (we will see why
a little later).
Another limitation of difference amplifiers is their relatively low input
impedance, typically in the range
of 10-500kW. That is not much of a
problem with a very low impedance
source such as our 100mW shunt, but
it becomes more of a concern as the
source impedance rises.
You can see from Fig.5 that any
source impedance will be in series
with the difference amplifier’s input
resistors R1a and R1b, potentially
impacting both the gain and CMRR.
A good rule of thumb is to make
sure the source impedance is lower
than the input impedance of the difference amplifier by the same order
of magnitude as the CMRR. So, for
a difference amplifier with a 90dB
CMRR and 10kW input resistors, the
source impedance should be less than
316mW. Any higher than that and the
CMRR will be adversely impacted.
Fig.5: this configuration delivers
a ground-referenced voltage
proportional to the difference
between the input voltages; Vout =
(R2 ÷ R1) × (Vin+ – Vin–).
In fact, the data sheets generally
specify CMRR with an input source
impedance of 0W. That is obviously
a totally unrealistic scenario – yet
another reason to be wary of data
sheet claims.
You might think that the CMRR
would be maintained if you had equal
source impedances on each input,
since both input resistors would be
increased by the same amount, but
no such luck. The manufacturer’s
laser-trimming matches the R1/R2
ratios in each divider, not necessarily their absolute values, which may
be a bit different.
Adding the same source resistance
to both inputs will likely unbalance
the ratios, making the CMRR worse.
By now you might be asking why
we should even bother with difference amplifiers if they have all these
limitations. Apart from the flexibility
we have already seen, and their role
in instrumentation amplifiers that
we will discuss soon, the difference
amplifier excels in the area of input
common-mode voltage range.
With the right resistor values, the
common-mode voltage can extend well
beyond the op amp’s power supply
rails. Off-the-shelf devices are readily
available with common mode input
ranges better than ±100V. I have built
discrete difference amps with common
Fig.4: eight possible ways to use the Fig.3 circuit to achieve different gains, level-shift signals and even sum/average
voltages.
Practical Electronics | March | 2025
27
Precision Electronics part three
Table 1: error budget for Fig.8 using an INA821
At Nominal 25°C
Error
Nominal Value
Shunt Resistor: RESI PCSR2512 (0.5%, 15ppm/˚C)
100mW
Differential Voltage due to I × Rshunt
100mV
Abs. Error
Rel. Error
Additional error over
0-50°C (Nominal ±25°C)
Abs. Error
0.50%
0.5mV
Rel. Error
0.038%
0.50%
0.0375mV
0.038%
InAmp: INA821 (Vos ±35µv, 5µV/˚C)
0mV
0.035mV
InAmp Input Voltage total (Line 2 + Line 3)
100mV
0.535mV
0.54%
0.1625mV
0.163%
InAmp Gain Resistor Rg: RN73C2A (0.1%, 10ppm/˚C)
2kW
2W
0.10%
0.5W
0.025%
InAmp Gain Error (0.015% ±35ppm/˚C)
0.125mV
0.02%
0.088%
InAmp Gain (Line 5 × Line 6)
25.7
0.0296
0.12%
0.0289
0.113%
Vout DM (Line 4 × Line 7)
2.57V
0.0167V
0.65%
0.0071V
0.275%
Vout CM (20V, 120db, ±1.5db over 0-50˚C)
0V
0.02mV
Vout (Line 8 + Line 9)
2.57V
0.0167V
mode voltages up to ±300V without
problems (but a lot of care).
Instrumentation amplifiers
One obvious solution to the difference amplifier’s input impedance
problem is to add a pair of unity-gain
input buffers in front of the input resistors, as shown in Fig.6. This solves
the input impedance problem (at the
expense of common mode voltage
range), but does nothing to help us
reach higher gains or achieve better
CMRR.
The classic three-op-amp instrumentation amplifier (or ‘inamp’) shown in
Fig.7 is a neat solution to the problem.
The two input op amps now work to
maintain the differential-mode voltage across resistor Rg. With this understanding, it is pretty easy to show
that this input stage has a differential
mode gain of Gdm = 1 + 2 × (R3 ÷ Rg)
Fig.6: this circuit
fixes the low input
impedance exhibited
by difference amplifiers
but it limits the input
voltage range and does
not add gain.
Fig.7: the classic threeop amp instrumentation
amplifier consists of a
high impedance gain
stage made up of two
op amps followed by
a difference amplifier.
This can provide
both higher gain and
improved CMRR
compared to difference
amplifier alone.
28
and a common mode gain of Gcm = 1.
We can see that with the right choice
of resistor values, this input stage can
improve the overall circuit’s differential gain but, given it has a common-
mode gain of one, it may not be as
obvious how this front-end can improve the overall CMRR.
Consider a situation where we want
an overall differential gain of 100 and
the highest possible CMRR. Imagine
the difference amplifier has a differential gain of 1 and a CMRR of 80dB.
The input stage will have a differential gain of 100 and a common-mode
gain of 1, giving a CMRR of 40dB.
The second stage adds 80dB of additional CMRR for a total circuit CMRR
of 120dB.
The instrumentation amp is effectively a gain stage with a CMRR equal
to the gain, followed by a common-
mode rejection stage with a differen-
0.0038mV
0.65%
0.0071V
0.275%
tial gain of unity or thereabouts. You
can now see why lots of the difference
amps on the market favour CMRR over
gain – they are intended for use in instrumentation amplifier applications.
Another nice feature of the instrumentation amp is that the gain can
be set by changing just one resistor,
Rg. This means practical devices can
have precision-trimmed matched resistors R1a/b, R2a/b and R3a/b, leaving the user to provide Rg externally
to set the gain. You can even switch
in different resistors to change the
gain or use a potentiometer to trim it.
A typical example of an off-theshelf instrumentation amplifier is the
INA821 from Texas Instruments (TI).
The data sheets show it has a CMRR
of 112dB for Gdm = 10 and 132dB for
Gdm = 100. This suggests they are getting 92dB of CMRR from the difference amp stage (and 20dB or 40dB
from the input stage).
The input impedance is 100GW,
which should be high enough for
pretty much any source impedance.
The cost of the INA821 is about $8.60
in single quantities, which is much
cheaper than anything you could
build yourself, given the very tighttolerance resistor matching required.
Let’s go through the process of designing the circuit of Fig.8 to compare with the ground-referenced circuit we built last time. We will build
up the error budget shown in Table
1 as we go.
Fig.9 shows the internal configuration of the INA821. We need a gain
of around 25, so we will choose Rg to
be 2kW, giving a gain of 25.7. The tolerance of this resistor is not critical
since we’ll trim the gain, but we do
care about its tempco. For this reason,
I chose the RN73C2A2K0BTD from
Practical Electronics | March | 2025
Difference & instrumentation amps
Table 2: error budget for Fig.8 using an LT1167A instead
At Nominal 25°C
Error
Nominal Value
Shunt Resistor: RESI PCSR2512 (0.5%, 15ppm/˚C)
100mW
Differential Voltage due to I × Rshunt
100mV
Abs. Error
Rel. Error
Additional error over
0-50°C (Nominal ±25°C)
Abs. Error
0.50%
0.5mV
Rel. Error
0.038%
0.50%
0.0375mV
0.038%
InAmp: LT1167A (Vos ±40µv, 0.2µV/˚C)
0mV
0.04mV
InAmp Input Voltage total (Line 2 + Line 3)
100mV
0.54mV
0.54%
0.0425mV
0.043%
InAmp Gain Resistor Rg: RN73C2A (0.1%, 10ppm/˚C)
2kW
2W
0.10%
0.5W
0.025%
InAmp Gain Error (0.02% ±10ppm/˚C)
0.005mV
0.02%
0.025%
InAmp Gain (Line 5 × Line 6)
25.7
0.0308
0.12%
0.0129
0.050%
Vout DM (Line 4 × Line 7)
2.57V
0.017V
0.66%
0.0024V
0.093%
Vout CM (20V, 106db over 0-50˚C)
0V
0.1002mV
Vout (Line 8 + Line 9)
2.57V
0.0171V
TE Connectivity. It has a tolerance
of 0.1% and a tempco of ±10ppm/°C.
The INA821’s input common mode
voltage range extends to within 2V
of either supply rail, so we need a
power supply of 22V or more on the
positive side and -2V or more on the
negative side.
I am going to assume we have a
+24V DC supply available, since this
would be the sort of input the power
supply’s series pass stage would need.
I have already used ±5V power rails
in my previous experiments, so I will
power the instrumentation amplifier
from +24V and –5V rails.
The INA821 has a maximum power
supply voltage of 36V, so this should
be fine, with a total of 29V applied
(24V + 5V). It is worth noting that it
is quite OK to power op amps asymmetrically like this, as long as you
understand that the input common
mode range and output swing will
likewise be asymmetrical.
We can now complete the error
budget table (Table 1). The first 8
lines of the table are similar to the
previous examples, arriving at a cumulative error of 0.65% with an ad-
Fig.8: an off-the shelf
instrumentation amplifier
(‘inamp’) can provide the
necessary gain (about 25) with
around 120dB of common-mode
rejection.
Practical Electronics | March | 2025
ditional 0.275% error over the 0°C to
50°C temperature range.
Unlike the previous circuit, we
now need to add the error due to the
common-mode signal making its way
through to the output. With a gain of
25.7, we can estimate the CMRR to
be 120dB based on 92dB for the difference amp stage plus 20log10(25.7)
= 28dB for the input stage.
With a common-mode voltage of
20V, we will therefore see 20µV at
the output. That’s insignificant compared to the 16mV of error due to the
differential-mode stage.
The change in CMRR with temperature is a bit harder to estimate.
TI provides graphs that show the
temperature variation of CMRR for
five sample devices at gains of one
and ten. From these, I have taken a
value ±1.5dB over 0°C to 50°C. It is
a bit of a guesstimation, but it does
not matter since the overall level of
common-mode feedthrough is so low
as to make this figure irrelevant.
The net result is shown therefore
shown at the bottom of Table 1. The
worst-case untrimmed error at 25°C
is ±0.65%, just a little worse than the
±0.55% error for the ground-referenced
circuit. In both cases, most of this error
is the 0.5% shunt resistor tolerance.
Unfortunately, the circuit’s perfor-
0V
0.66%
0.0024V
0.093%
mance over the temperature range
is not great. We are seeing ±0.275%
error, with two major contributors:
the instrumentation amplifier’s input
offset voltage drift and its gain drift.
The LTC2057-based circuit was much
better at 0.075%, as we would expect
from an auto-zero op amp.
Doing better – but at a price
I wanted to see if we could improve
on this, so I looked for a ‘better’ instrumentation amp. The LT1167A
fits the bill. Its input offset voltage
at 25°C is similar to the INA821, but
its offset drift is 25 times better at
0.2µV/°C. Its gain drift with temperature is also better at ±10ppm°/C, compared with the ±35ppm/°C. Table 2
shows the error budget for this version of the circuit.
As an aside, it’s a good idea to create
these error budget tables in a program
like Excel or LibreOffice Calc. I set up
the formulas so that I can easily try
new parts and have the whole table
recalculate automatically.
Compared to the INA821, the new
circuit shows a similar error at nominal
temperature of ±0.66%, but an error
over the temperature range three times
better at 0.093%. So, we should use
this device, right? Well, the LT1167A
costs $30 each in one-off quantities,
Fig.9: the
INA821 has six
laser-trimmed
precision
resistors and
three op amps.
The user must
provide an
external resistor
(Rg) to set the
overall gain.
29
Precision Electronics part three
Fig.10: the measured untrimmed data for the INA821based circuit shows about 0.3% gain error; most of this is
due to the shunt resistor tolerance.
so we would want to be certain there
was no alternative.
It should however come as no surprise that precision components that
are at the very extremes of performance
will be costly. The manufacturers know
full well that if there are no or few alternatives, you will have to pay up.
Test results
I spared no expense and tested both
devices. I built the circuits and measured the input current vs output voltage characteristics with both zero and
the full common-mode voltage of 20V.
The results for VCM = 20V are shown
in Tables 3 and 4, and plotted in Figs.10
& 11.
For the INA821, the untrimmed
errors range from 0.01% at zero current to around 0.33% at 1A. The results were a little better with zero
common mode voltage. As expected,
this is better than our error budget’s
0.65% worst-case estimate. The errors
increase steadily with the magnitude
Fig.11: the untrimmed data for the LT1167A-based circuit
shows the same 0.3% gain error as Fig.10 but has more offset
error. It should perform better over the temperature range.
of current, suggesting a gain error is
the main contributor.
The graphed results and line of best
fit shows this to be the case. The offset
correction we need to apply is very low
(around 250µV) and the gain error is
about 0.3% (the measured gain is about
0.3% higher than we expect). Again,
the shunt resistor with its 0.5% tolerance is likely to be the culprit.
After correcting the results, we get
a trimmed error of ±0.03%, very comparable with the ground-referenced
circuit. However, our concern with
the INA821 circuit is its performance
over temperature.
The measured CMRR of this circuit
was 106dB – not as good as the estimates of 120dB, but nevertheless acceptable. It’s actually a bit difficult to
measure CMRR, since things like op
amp input offset voltage can also change
over the common-mode range, and it’s
impossible to isolate the causes with a
simple output voltage measurement.
The LT1167A circuit has worse un-
Table 5: theoretical improvement to Table 1 with dynamic zero trim
trimmed accuracy, peaking at almost
0.47%, but again the graphs show it
to be almost all gain error. After trimming, the error is reduced to ±0.025%,
very similar to the INA821. The temperature coefficient is better, of course.
Another solution
Rather than commit to a $30 chip,
I want to introduce another trick we
can use to improve precision in this
type of situation.
So far, we have applied fixed offset
and gain corrections to minimise the
static errors in the circuit. In practice,
this would be done for each sample in
software, based on some one-off calibration performed at a standard temperature when we initially set up the
instrument (and maybe when we periodically re-calibrated it).
Another approach might be to try
to obtain the corrections in real-time
at the ambient operating temperature.
High-end instruments, like the 6½ digit
multimeters that I used to obtain the
At Nominal 25°C
Abs. Error
Rel. Error
Additional error over
0-50°C (Nominal ±25°C)
Error
Nominal Value
Shunt Resistor: RESI PCSR2512 (0.5%, 15ppm/˚C)
100mW
Abs. Error
Differential Voltage due to I × Rshunt
100mV
0.5mV
InAmp: INA821 (Vos ±35µv, 5µV/˚C) – zero trimmed
0mV
0.035mV
InAmp Input Voltage total (Line 2 + Line 3)
100mV
0.535mV
0.54%
0.0375mV
0.038%
InAmpGain Resistor Rg: RN73C2A (0.1%, 10ppm/˚C)
2kW
2W
0.10%
0.5W
0.025%
0.50%
InAmp Gain Error (0.015% ±35ppm/˚C)
0.50%
Rel. Error
0.038%
0.0375mV
0.038%
0mV
0.02%
0.088%
InAmp Gain (Line 5 × Line 6)
25.7
0.0296
0.12%
0.0289
0.113%
Vout DM (Line 4 × Line 7)
2.57V
0.0167V
0.65%
0.0039V
0.150%
Vout CM (20V, 120db, ±1.5db over 0-50˚C)
0V
0.02mV
Vout (Line 8 + Line 9)
2.57V
0.0167V
30
0.0038mV
0.65%
0.0039V
0.150%
Practical Electronics | March | 2025
Difference & instrumentation amps
Measured Data
Fig.12: we can improve the
temperature-dependent error of
the circuit by adding switches to
dynamically measure the offset, like
an auto-zero op amp.
results shown here, effectively perform
a zero and full-scale calibration every
20ms measurement cycle. Any temperature drift is calibrated out moreor-less in real-time.
We are not aiming for anything near
that level of precision, but a simpler
version can be a useful technique. It
is pretty difficult to do a full-scale
calibration of our test circuit, as we
would need a precision 1A current
source, but we could do a zero calibration fairly easily.
This won’t let us trim out gain drift
due to temperature but would let us
calibrate out temperature-dependent
offset errors in real-time – a bit like
auto-zero op amps do.
Let’s take a look at this approach
using the INA821 example. Looking
at the error budget table, we can see in
line 3 that we have a possible ±125µV
drift in offset voltage over the temperature range. If we could calibrate that
out, as shown in Table 5, we would
almost halve the temperature error
from ±0.275% to ±0.15%.
Fig.12 shows one way we could
achieve this in practice. Normally, S1
would be closed and S2 open so that
we could take current measurements
as before. Opening S1 and closing S2
shorts the inputs of the instrumentation amplifier so that we can use the
ADC to read the circuit’s offset voltage.
We would still need a fixed gain correction as before, but we can use the
zero-scale reading to create a dynamic offset correction that will eliminate
some of the temperature drift error.
Extending the range
Let’s regroup and consider what we
have achieved so far.
Practical Electronics | March | 2025
Untrimmed Error
Trimmed Error
I (mA)
Vout (mV)
Absolute (mV)
Relative
Absolute (mV)
Relative
0.000
0.233
0.23
0.01%
0.48
0.019%
99.726
256.654
0.36
0.01%
-0.19
-0.007%
199.824
514.948
1.40
0.05%
0.05
0.002%
299.980
772.739
1.79
0.07%
-0.36
-0.014%
400.008
1031.164
3.14
0.12%
0.19
0.008%
499.980
1289.040
4.09
0.16%
0.34
0.013%
600.007
1546.980
4.96
0.19%
0.41
0.016%
699.965
1804.750
5.84
0.23%
0.49
0.019%
800.024
2062.770
6.71
0.26%
0.56
0.022%
899.971
2320.490
7.56
0.29%
0.61
0.024%
999.866
2578.110
8.45
0.33%
0.70
0.027%
Table 3 – untrimmed measured results from the INA821 circuit shown in Fig.8.
Measured Data
Untrimmed Error
Trimmed Error
I (mA)
Vout (mV)
Absolute (mV)
Relative
Absolute (mV)
Relative
0.000
1.614
1.61
0.06%
0.46
0.018%
99.759
258.745
2.37
0.09%
0.14
0.005%
199.898
517.182
3.44
0.13%
0.14
0.005%
299.829
775.181
4.62
0.18%
0.24
0.009%
400.044
1033.716
5.60
0.22%
0.15
0.006%
500.013
1291.840
6.81
0.26%
0.28
0.011%
600.390
1549.970
6.97
0.27%
-0.64
-0.025%
700.009
1807.980
8.96
0.35%
0.28
0.011%
800.060
2066.110
9.96
0.39%
0.20
0.008%
899.975
2323.980
11.04
0.43%
0.21
0.008%
999.872
2581.780
12.11
0.47%
0.20
0.008%
Table 4 – untrimmed measured results from the INA821 circuit shown in Fig.8
when replaced with an LT1167A.
We have shown that with the shunt
in the positive supply, we can probably achieve a trimmed accuracy of
around 0.03% at 25°C with an additional 0.15% error over the 0–50°C temperature range if we use the INA821
instrumentation amplifier and dynamic
offset correction. Let’s call this 0.2%
of total error.
This suggests we will have an overall resolution of ±2mA in our 1A current (ignoring ADC precision for now).
That is not good enough to measure
the microamp resolution we would
like to achieve.
I hope it is clear by now that we
are not going to get the required three
orders of magnitude improvement
in precision just by improving the
signal conditioning circuit. Even if we
could, we will run into ADC quantisation limits, which we will cover in
a later article.
The current circuit needs an ADC
with at least 10 effective bits of resolution – three more orders of magnitude
would require over 33 bits of effective
resolution, which is pushing the limits
of what is possible!
There is another way. We could
pretty easily scale the range of the circuit by using a different shunt resistor. For example, using a 10W resistor
would give a range of 0 to 10mA with
±20µA resolution, while a 1kW resistor would yield a range of 100µA fullscale with ±200nA resolution.
That will require some additional
circuitry to switch the ranges. This,
and the dynamic offset zeroing, will
require us to add some switching elements to our signal path, which will
themselves introduce some imprecision. We will look more deeply into
signal switching in the next instalment
of this series.
PE
31
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